Two-wire dimmer with improved zero-cross detection

ABSTRACT

A lighting device, such as a two-wire lighting control device, may include a controllably conductive device and a control circuit. The controllably conductive device may supply an AC line voltage to a load in response to a dive signal such that the controllable conductive device is non-conductive for a first duration of time and conductive for a second duration of time within a half-cycle of the AC line voltage. The control circuit may receive a signal from the controllably conductive device that represents a voltage developed across the controllable conductive device during the first duration of time. The control circuit may generate a sine-wave-shaped signal that complements the voltage developed across the controllably conductive device during the second duration of time. The control circuit may also filter the signal from the controllably conductive device during the first duration of time and the sine-wave-shaped signal during the second duration of time.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.14/839,443, filed Aug. 28, 2015, which is a continuation of U.S. patentapplication Ser. No. 13/793,245, filed Mar. 11, 2013, which issued asU.S. Pat. No. 9,155,162 on Oct. 6, 2015, and which claims the benefit ofU.S. Provisional Application No. 61/700,964, filed Sep. 14, 2012, all ofwhich are incorporated by reference in their entirety.

BACKGROUND

Load control devices and systems control the amount of power deliveredfrom an alternating-current (AC) power source to an electrical load,such as a lighting load, for example. Such lighting control systemstypically employ a controllably conductive device, such as a thyristoror triac for example, for controlling the intensity of the lightingload. The controllably conductive device is rendered conductive at aphase angle during each half-cycle of the AC power source in response toa trigger signal received at a control input. This establishes, withineach half-cycle, a conduction period where power is being delivered tothe load and a non-conduction period where power is not being deliveredto the load.

In a typical forward phase-control system, generation of the triggersignal is synchronized with the AC line voltage. At some time after azero-crossing of the AC line voltage is detected, the trigger signal isgenerated, and the controllably conductive device is renderedconductive. The controllable conductive device remains conductive forthe remainder of the AC half cycle. During the time interval between thedetection of the zero-crossing and the generation of the trigger signal,the controllable conductive device is non-conductive. This time intervalmay also be referred to as the phase or firing angle of the system. Byvarying this time interval, the effective power delivered to the load isvaried. Typically, this time interval is altered in response toadjustment of a dimming knob or slider by a user and/or in response tochanges in a dimming signal level.

FIGS. 1A-1D depict example AC voltage waveforms as measured across thecontrollably conductive device. When the controllably conductive deviceis non-conductive, the complete AC voltage waveform, as shown in FIG.1A, is developed across the device. At a relatively low light level, asshown in FIG. 1B, the controllably conductive device is non-conductivefor a first duration 102 of the half-cycle (i.e., from the zero-crossingof the current half-cycle of the AC voltage waveform to a point withinthe half-cycle). The trigger signal is generated (shown as point “A”).Then, the controllably conductive device is conductive for a secondduration 104 of the half-cycle. FIGS. 1C and 1D illustrate waveforms ata 50% dimming level and a relatively high light level, respectively.

At low levels of delivered power, like that depicted in FIG. 1B, even asmall variation in the phase angle (and thus the conduction period)usually represents a relatively large variation in the percentage of thetotal RMS power delivered to the load. At these low power levels, anyvariation of the phase angle, whether between AC cycles or over periodsof time, can be manifested as annoying and unacceptable intensitychanges, including visible flickering of the light source. Since thephase angle is dependent on the detection of the zero-crossing, it iscrucial that zero-cross detection be accurate and reliable. AC lineconditions, however, are rarely ideal. And, less than ideal conditionscan cause inaccuracy in the detection of zero-crossings, with consequentintensity variations and/or flickering, as well as other problems,especially at low levels of delivered power. One condition that cancause intensity variations and/or flickering is intermittent and/orperiodic electrical noise on the AC line.

FIGS. 2A-2C illustrate example AC voltage waveforms having noise. Forexample, as illustrated in FIG. 2A, voltage spikes can be imposed on anAC line, which may occur when heavy equipment, such as large motorloads, are switched on and off. Electrical noise on an AC line, such asthese spikes, may be incorrectly interpreted by dimming circuitry as oneor more zero-crossings of the AC line voltage. Such falseinterpretations can lead to erratic intensity variations and/orflickering in the lighting load. Another common characteristic ofelectrical noise on an AC line may include a bumpy or wavy distortion,as shown in FIG. 2B, which can also cause false zero-crossing detection.The presence of harmonics of the AC fundamental on the AC line isanother condition that can cause false zero-crossing detection. Thepresence of harmonics may change the shape of the AC line voltagewaveform from a pure sinusoid to a generally sinusoidal waveform, havingflattened peaks rather than round peaks, as illustrated in FIG. 2C.

One approach to mitigate the effects of noise on an AC line includesfiltering the AC line voltage prior to performing zero-crossingdetection. For example, the Real-Time Illumination Stability System(RTISS) uses a filter to improve the performance of a dimming system.The RTISS technology is described in commonly-assigned U.S. Pat. No.6,091,205, issued Jul. 18, 2000, and U.S. Pat. No. 6,380,692, issuedApr. 30, 2002, both entitled Phase controlled dimming system with activefilter for preventing flickering and undesired intensity changes, theentire disclosures of which are hereby incorporated by reference.

Both three-wire dimming systems and two-wire dimming systems may employthe RTISS technology. FIG. 3A depicts an example three-wire dimmingsystem 300. FIG. 3B depicts an example two-wire dimming system 302. Bothdimming systems have dimmer switches 304, 306 electrically coupledbetween an AC power source 308 and an electrical load 310. The dimmerswitches 304, 306 are connected to the AC power source 308 by a firstwire 312 (also referred to as a “hot” wire) and to the load 310 by asecond wire 314 (also referred to as a “dimmed-hot” wire). However, thethree-wire dimmer switch 304 also has a third wire 316 (also referred toas a “neutral” wire), which provides a path back to the return side ofthe AC power source 308. The two-wire dimmer switch 306 is not connectedto the neutral wire 316.

The three-wire dimmer switch 304 has two waveforms available to it. Afull (i.e., not switched) AC line voltage waveform 318 is available tothe three-wire dimmer 304, by virtue of its third wire 316. Adimmer-voltage waveform 320, measured from the first wire 312 and thesecond wire 314, is also available to the three-wire dimmer 304. Thethree-wire dimmer switch 304 is able to use the full AC line voltagewaveform 318 for filtering to determine the zero-crossings of the ACline voltage waveform of the AC power source 308 and to generate an ACload voltage waveform 322 (e.g., a dimmed-hot voltage that is measuredfrom the second wire 314 and the third wire 316). The two-wire dimmerswitch 306, on the other hand, without a path back to return side of theAC power source 308, only has the dimmer-voltage waveform 320 at itsdisposal, and not the full AC line voltage waveform 318.

In two-wire dimming systems, variations in phase delay associated withfiltering (e.g., from the input to output of the filter) may affect thestability of the dimming system and/or the amount of error in thezero-crossing detection. Having only the dimmer-voltage waveform 320available for filtering to determine the zero-crossings of the AC linevoltage, the two-wire dimmer switch 306 may experience substantialvariation in phase delay through the filter as a function of the firingangle of the controllably conductive device. To illustrate, FIG. 4provides a plot 402 that shows how phase delay through a low-pass filtermay vary as a function of firing angle in a two-wire dimmer switch. Forexample, as shown, a relatively large firing angle (i.e., a relativelysmall conduction period) may correspond to a relatively large phasedelay. As the firing angle decreases (for example, from approximately 7milliseconds to 2 milliseconds, as shown) and the conduction periodincreases, the phase delay decreases substantially (for example, fromapproximately 5.5 milliseconds to 3 milliseconds).

The variation in phase delay may affect system stability and/or theamount of error in the zero-crossing detection. Errors in zero-crossdetections may further exacerbate the phase delay problem through thefilter, which in turn may further increase the errors in subsequentzero-crossing detections. This positive feedback effect may lead tosystem instability, in the form of a runaway condition, for example.

SUMMARY

As disclosed herein, a lighting device, such as a two-wire lightingcontrol device, may include a controllably conductive device and acontrol circuit. The controllably conductive device may supply an ACline voltage to a load, being non-conductive for a first duration oftime and conductive for a second duration of time within a half-cycle ofthe AC line voltage. The control circuit may receive a signal from thecontrollably conductive device that represents a voltage developedacross the controllable conductive device during the first duration oftime. The control circuit may generate a sine-wave-shaped signal thatcomplements the voltage developed across the controllably conductivedevice during the second duration of time. For example, the sine-waveshaped signal may be representative of the magnitude of the voltagegenerated across the load during the second duration of time. Thecontrol circuit may also filter the signal from the controllablyconductive device during the first duration of time and thesine-wave-shaped signal during the second duration of time.

BRIEF DESCRIPTION OF THE DRAWINGS

FIGS. 1A-D depict example alternating current (AC) voltage waveformsdeveloped across a controllably conductive device in a forwardphase-control dimming system.

FIGS. 2A-C illustrate example AC voltage waveforms having noise.

FIGS. 3A and 3B illustrated an example three-wire dimming system and atwo wire dimming system, respectively.

FIG. 4 is a plot showing the relationship between phase delay through afilter and firing angle in a two-wire dimming system.

FIG. 5 is a functional block diagram of an example two-wire dimmingsystem.

FIGS. 6A-6E illustrate various combined signals, with example non zeromagnitude signal waveforms.

FIG. 7 is a block diagram of a portion of a two-wire lighting controldevice having a digital-to-analog converter for generating anon-zero-magnitude signal.

FIG. 8 is a simplified flowchart of a zero-crossing procedure executedperiodically by a control circuit of the lighting control device of FIG.7.

FIG. 9 is a simplified flowchart of a dimming timer procedure executedby the control circuit of the lighting control device of FIG. 7.

FIG. 10A is a block diagram of a portion of a two-wire lighting controldevice that uses pulse-width modulation to generate a non-zero-magnitudesignal; FIGS. 10B and 10C present corresponding signal diagrams.

FIG. 11 is a simplified flowchart of a bin set procedure executedperiodically by a control circuit of the lighting control device of FIG.10A for generating the non zero-magnitude signal.

FIG. 12 is a simplified flowchart of a bin reset procedure executed bythe control circuit of the lighting control device of FIG. 10A forgenerating the non zero-magnitude signal.

FIG. 13 is a simplified schematic diagram of a portion of an exampletwo-wire lighting control device.

FIG. 14 is a simplified schematic diagram of a sampling procedureexecuted periodically by a control circuit of the lighting controldevice of FIG. 13.

DETAILED DESCRIPTION

A two-wire lighting control device, may mitigate variations in phasedelay through a filter by generation and application of anon-zero-magnitude signal (e.g., a fill signal) to the input of thefilter. To illustrate, FIG. 5 is a functional block diagram of anexample two-wire dimming system 500. The system 500 may include atwo-wire lighting control device 502 (e.g., a dimmer switch or a dimmingunit) connected in series with an AC power source 504 and an electricalload 506, such as an illumination load.

The two-wire lighting control device 502 may include a controllablyconductive device 508, such as a thyristor, for example, a triac,electrically coupled in series between the AC power source 504 and theelectrical load 506. The controllable conductive device 508 mayalternatively comprise a field-effect transistor (FET) in a rectifierbridge, two FETs in anti-series connection, one or more insulated-gatebipolar junction transistors (IGBTs), back-to-back silicon-controlledrectifiers (SCRs), or any suitable bidirectional semiconductor switch.

The controllably conductive device 508 is controlled by a controlcircuit 510 (e.g., a controller) for causing the controllably conductivedevice 508 to be conductive or non-conductive. The control circuit 510may control the controllably conductive device 508 directly or through adrive circuit 512, as illustrated in FIG. 5. The drive circuit 512 mayinclude circuitry to convert control signals from the control circuit510 to signals suitable for rendering the controllably conductive device508 conductive or non-conductive. The timing of the controllablyconductive device 508 becoming conductive and non-conductive may beestablished by the control circuit 510 and set to establish a desireddimming level in the load 506. The control circuit 510 may set thetiming of the controllably conductive device 508 in accordance with theoperation of a computer program and/or as manually set by the user, forexample.

The control circuit 510 may include a microprocessor, a microcontroller,a programmable logic device (PLD), an application specific integratedcircuit (ASIC), a field programmable gate array (FPGA), or any suitablecontrol circuit. The control circuit 510 may include hardware, firmware,software, and/or a combination of the hardware, firmware, and/orsoftware suitable for controlling the controllably conductive device508. The control circuit 510 may include and/or interface with a memory514 (i.e., memory may be internal or external to the control circuit).The memory 514 may include any component suitable for storinginformation. For example, the memory 514 may include volatile and/ornon-volatile memory. The control circuit 510 may include programmaticaspects, such as stored computer instructions, and/or storable dataassociated with the operation of the two-wire lighting control device502. These programmatic aspects may be stored in and/or retrieved frommemory 514.

The control circuit 510 may interface with input/output devices such asa communication circuit 516 and/or a user interface 518. Thecommunication circuit 516 may include any component suitable for thetransmission and reception of data, such as a modem, for example. Theuser interface 518 may include any component suitable for presenting andreceiving information from a user. For example, the user interface 518may include one or more buttons, switches, sliders, or the like. Theuser interface 518 may include a display, such as one or morelight-emitting diode (LED) indicators, a liquid crystal display (LCD)screen, a touch screen display, or the like. The control circuit 510 mayreceive commands, configuration information, or the like, via thecommunication circuit 516 and/or the user interface 518. The controlcircuit 510 may transmit information (such as the present dimming levelfor example, commands, or the like) via the communication circuit 516and/or the user interface 518. For example, the control circuit 510 may,via the user interface 518, receive a specific desired dimming levelfrom a user and confirm the user's input. Also for example, the controlcircuit 510 may receive a command from another device via thecommunication circuit 516 and/or send a command to another device. Thetwo-wire lighting control device 502 also comprises a power supply 519coupled in parallel with the controllably conductive device 508 forconducting a charging current through the load 506 to generate a DCsupply voltage for powering the control circuit 510 and otherlow-voltage circuitry of the lighting control device.

A first signal 520 may be associated with the controllably conductivedevice. For example, the first signal 520 may represent thedimmer-voltage waveform 320 developed across the controllably conductivedevice 508, as depicted in FIG. 5. The first signal 520 may betransformed to make it more suitable for processing. For example, thefirst signal 520 may represent the dimmer-voltage waveform across thecontrollably conductive device with a DC component added, such that thefirst signal 520 maintains a positive magnitude. With regard to thedimming level, the first signal 520 may be consistent with the signalsillustrated in FIGS. 1B-1D. For example, the first signal 520 may have afirst portion for a first duration of time of the AC half-cycleassociated with the controllably conductive device 508 beingnon-conductive, and a second portion for a second duration of time ofthe AC half-cycle associated with the controllably conductive device 508being conductive. To illustrate, in an example operation, after azero-crossing of the AC line voltage is detected, the control circuit510 waits for the first duration, which may be commensurate with thedesired dimming level, before generating the trigger signal. The triggersignal controls the controllably conductive device 508 to change fromnon-conductive to conductive, after which the first signal 520 mayeffectively represent zero volts until the end of the AC half cycle(i.e., consistent with a forward phase-control signal). Of course, thefirst signal 520 may alternatively be consistent with a reversephase-control operation or a center phase-control operation of thecontrollably conductive device 508.

A signal generator 522 may generate a non-zero-magnitude signal 524 andmay comprise any circuit suitable for generating a signal. For example,the signal generator may be incorporated into and/or may usefunctionality of the control circuit 510. For example, the signalgenerator 522 may include a digital signal being generated by thecontrol circuit and output via a digital-to-analog converter. Also forexample, the signal generator 522 may generate one or more pulse-widthmodulated signals from the control circuit 510 and convert those one ormore pulse-width modulated signals into a step-sine wave.

The non-zero-magnitude signal 524 may be generated based on theoperation of the control circuit 510. For example, thenon-zero-magnitude signal 524 may be generated such that it complementsthe first signal 520 from the controllable conductive device. Forexample, the non-zero-magnitude signal 524 may be a fill signal,relative to the first signal 520 from the controllably conductivedevice, in that it fills in the “missing portion” of the first signal520 (e.g., the portion of the first signal that represents effectivelyzero volts, such as a first signal with an effectively zero amplitude,for example). The non-zero-magnitude signal 524 may be sine-wave-shaped.For example, the non-zero signal may be a step-sine wave that includessteps based on at least one pulse-width modulated signal, such as apulse width modulated signal from the control circuit 510. Thestep-sine-wave may also include steps based on at least one phasecorrection corresponding to a zero-crossing detected by thezero-crossing detector.

The first signal 520 and the non-zero-magnitude signal 524 may becombined into a combined signal 528, which is passed through a filtercircuit 526. The non-zero-magnitude signal 524 may be shaped such thatthe filter delay associated with filtering the first signal 520 and thenon-zero-magnitude signal 524 is consistent with the filter delayassociated with filtering the AC line voltage. In effect, inclusion ofthe non-zero-magnitude signal 524 may mitigate delay variation, as afunction of the dimming level, through the filter circuit 526. Thus, thenon-zero-magnitude signal 524 may be shaped to maintain stability of thetwo-wire lighting control device 502.

The first signal 520 and the non-zero-magnitude signal 524 may be addedtogether to form the combined signal 528 for the filter circuit 526. Inthis example, the non-zero-magnitude signal 524 may have a magnitude ofeffectively zero volts when the first signal 520 has a non-zeromagnitude, and the non-zero-magnitude signal 524 may have a non-zeromagnitude when the magnitude of the first signal 520 is effectively zerovolts.

The first signal 520 and the non-zero-magnitude signal 524 may becombined together by a combiner circuit 525 to form the combined signal528 for the filter circuit 526. In this example, the two-wire lightingcontrol device 502 may include a switch (not shown), such as amultiplexer for example. The multiplexer may receive the first signal520 on a first input and may receive the non-zero-magnitude signal 524on a second input. The multiplexer may select the first input or thesecond input to output in dependence upon a select signal. The selectsignal may be based on the inputs. For example, when the first signal520 has a non-zero magnitude, the multiplexer may select the first input(i.e., the first signal 520) to output, and when the magnitude of thefirst signal 520 is effectively zero volts, the multiplexer may selectthe second input (i.e., the non-zero-magnitude signal) to output. Theselect signal may be based on the trigger signal generated at thecontrol input of the controllably conductive device 508 by the controlcircuit 510. For example, the multiplexer may operate in concert withthe trigger signal between the control circuit 510 and the controllablyconductive device 508. The multiplexer may select the second input(i.e., the non-zero-magnitude signal 524) when the control signalbetween the control circuit 510 and the controllably conductive device508 directs the controllably conductive device 508 to become conductive(i.e., at the phase angle). And, the switch may select the first input(i.e., the first signal 520) when the controllably conductive device 508becomes non-conductive at the end of the half-cycle.

To illustrate, the controllably conductive device 508 may benon-conductive for a first duration of time and conductive for a secondduration of time. The multiplexer may select the first signal 520 duringthe first duration of time and the non-zero-magnitude signal 524 duringthe second duration of time. And, as illustrated in FIG. 5, the combinedsignal 528 may include a portion of the first signal 520 (shown in solidline) and a portion of the non-zero-magnitude signal 524 (shown inbroken line). In the resultant combined signal 528, thenon-zero-magnitude signal portion may serve to complement (e.g., tocomplete) the first signal portion. In other words, thenon-zero-magnitude signal 524 may be a fill-signal that fills-in amissing portion of the first signal 520.

The filter circuit 526 may filter the combined signal 528 to attenuatehigh-frequency components, particularly high-frequency noise associatedwith the first signal 520 that may tend to cause errors in zero-crossingdetection. The filter circuit 526 may be consistent with the filtersdisclosed in previously-referenced U.S. Pat. Nos. 6,091,205 and6,380,692. For example, the filter circuit 526 may be a low-pass filter,such as, for example, a Bessel filter for example, which may beconfigured to provide a substantially linear phase delay of less thanone-half of a period of the fundamental frequency. The low-pass filtermay be configured to attenuate frequency components of the combinedsignal 528 that are substantially equal to third order harmonics andgreater of the fundamental frequency.

The filtered output is received by a zero-cross detection circuit 530,which may detect when the magnitude of the filtered output crosseseffectively zero volts (i.e., goes from positive to negative and/or fromnegative to positive). The zero-cross detection circuit 530 may provideinformation indicative of the timing of the zero-crossings to thecontrol circuit 510. And, as described above, the control circuit 510causes the controllably conductive device 508 to become conductive ornon-conductive with timing in accordance with a desired dimming level inthe load 506. For example, the timing between a zero-crossing and thesubsequent trigger signal to render the controllable conductive deviceconductive may be commensurate with the desired dimming level.

FIGS. 6A-6E illustrate various combined signals, along with examplenon-zero-magnitude signal waveforms. The shape or waveform of thenon-zero-magnitude signal 524 generated by the signal generator 522 maygenerally complement the first signal 520 to maintain stability of theoverall system. For each example shown in FIGS. 6A-6E, the portion ofthe combined signal 528 associated with the first signal 520 is shown insolid line, and the portion of the combined signal associated with thenon-zero-magnitude signal 524 is shown in broken line. Of course, one ofordinary skill will appreciate that the signals represented are examplesand may be adapted for use in forward phase-control, reversephase-control, and/or center phase-control systems.

As shown in FIG. 6A, the combined signal may represent a smoothsinusoid. Signal 602 is an example combined signal associated with acontrollably conductive device 508 operating with a low-end firingangle. Signal 604 is an example combined signal associated with acontrollably conductive device operating at a 50% firing angle. And,signal 606 is an example combined signal associated with a controllablyconductive device 508 operating at a high-end firing angle. Thesinusoidal combined signal shown in FIG. 6A maintains a fundamentalfrequency consistent with that of the AC line voltage of the AC powersource 504. As a result, variation in the phase delay through the filtercircuit 526 as a function of the firing time may be mitigated.

Different waveform shapes may be suitable for the portion of thecombined signal associated with the non-zero-magnitude signal 524. FIG.6B illustrates a non-zero-magnitude signal that comprises a square wavehaving constant amplitude. Signals 608, 610, and 612 represent low-end,50%, and high-end, respectively. Here, the pulse width of the squarewave varies with the firing angle; for example, as shown in FIG. 6B, thesquare wave is narrow at low-end and is much wider at high-end.

FIG. 6C shows that the width and the amplitude of the square wave may beadjusted according to the firing angle. Signals 614, 616, and 618represent low-end, 0%, and high-end respectively. For example, theamplitude of the square wave may be selected to match that of the firstsignal at the firing time. Thus, at low-end, the square-wave fill signalis narrow and has a relatively of low amplitude. At 50%, the square-wavefill signal has a width of one-half of the half-cycle of the AC inputsignal and is at its highest amplitude (e.g., the peak amplitude of theAC input signal). At high-end, the square-wave fill signal may be at itswidest, having a low amplitude, similar to that at low-end.

FIG. 6D shows a non-zero-magnitude signal that may comprise a triangularwave. Signals 620, 622, and 624 represent low-end, 50%, and high-endrespectively. At low-end and at 50%, the fill signal may have an initialvoltage similar to that of the first signal and then steadily decreaseto zero. At high-end, the triangular waveform may be shaped to roughlyapproximate the sinusoid of the AC line voltage of the AC power source504. For example, the non-zero-magnitude signal may extend, increasingthe amplitude to a peak approximately at the center of the half-cycle ofthe AC line voltage and then extend, decreasing to zero volts at oraround the zero-crossing of the AC power source 504.

FIG. 6E shows that a step-sine wave may be used as a non-zero-magnitudesignal. Here, the step-sine wave may include a composite of multiplesquare waves of varying width and amplitude to approximate a sine wave.Signals 626, 628, and 630 represent low-end, 50%, and high-endrespectively.

Each of the example combined signals maintains a fundamental frequencyconsistent with that of the original AC input signal such that variationin the phase delay through the filter as a function of the firing timemay be mitigated. In effect, the non-zero-magnitude signals may providea corrective influence when the resultant combined signal is filtered.For example, when the first signal is mostly a sinusoid and a smallerportion of the first signal is effectively zero volts, the added areafrom an example fill signal provides a correction to shift the centerfundamental of the filter output and to compensate for any error in thefiring angle. Similarly, when the first signal is represented by a smallportion of sinusoid and a larger portion of zero, then the fill signalprovides additional negative feedback to correct the shift in center ofthe fundamental of the filter output and to correct the correspondingerror in the firing angle.

The example combined signals, shown in FIGS. 6A-E may be used forpurposes alternative to or in addition to serving as input to a filter.For example, a combined signal may be used to make a power calculation,as disclosed in commonly-assigned U.S. patent application Ser. No.13/793,308, entitled POWER MEASUREMENT IN A TWO-WIRE LOAD CONTROLDEVICE, filed Mar. 11, 2013, and patented as U.S. Pat. No. 9,250,669 onFeb. 2, 2016, which is hereby incorporated by reference. For example,the combined signals shown in FIGS. 6A and 6B, may accurately depict anAC voltage waveform such that a useful power calculation may be made.

FIG. 7 is a block diagram of a portion of a two-wire lighting controldevice 700. The lighting control device 700 comprises a hot terminal Hadapted to be coupled an AC power source (not shown) and a dimmed-hotterminal DH adapted to be coupled to a lighting load (not shown). Thelighting control device 700 comprises a controllably conductive devicethat is implemented as two FETs 710, 712 coupled in anti-seriesconnection. The gates of the FETs 710, 712 are coupled to a gate drivecircuit 714 and the sources of the FETs are coupled together at circuitcommon. A control circuit 720 is coupled to the gate drive circuit 714and generates a drive signal V_(DR) for controlling the gate drivecircuit to render the FETs 710, 712 conductive and non-conductive tothus control the amount of power delivered to the lighting load.

The lighting control device 700 comprises a power supply (not shown)that may be coupled across the controllably conductive device forgenerating a supply voltage V_(CC) (e.g., approximately 3.3 volts) forpowering the control circuit 720. The lighting control device 700 alsocomprises a reference supply (not shown) for generating a referencevoltage V_(REF), which may have a magnitude equal to approximatelyone-half of the DC supply voltage V_(CC) (e.g., approximately 1.5volts). The reference supply may comprise, for example, a simple powersupply, such as a resistor-zener power supply.

The lighting control device 700 generates a combined signal 730 from adimmer-voltage signal 732 and a non-zero-magnitude signal 734. Thelighting control device 700 comprises a scaling circuit 730 coupledacross the controllably conductive device (i.e., theanti-series-combination of the FETs 710, 712) for producing a scaledversion of the dimmer-voltage waveform developed across the controllablyconductive device to which the reference voltage V_(REF) is added togenerate the dimmer-voltage signal 732. Accordingly, the dimmer-voltagesignal 732 is representative of the dimmer-voltage waveform developedacross the controllably conductive device.

The lighting control device 700 comprises a digital-to-analog converter(DAC) 742 coupled to the control circuit 720 for generating thenon-zero-magnitude signal 734. In the example of FIG. 7, the controlcircuit 720 may generate the non-zero-magnitude signal 734 by sendingdigital signal values to the digital-to-analog converter 742. Forexample, the control circuit 720 may have a look-up table of valuesstored in memory that may represent samples of the non-zero-magnitudesignal 734. For example, the values may represent samples of any of thenon-zero-magnitude signals, as shown in FIGS. 6A-6E. The control circuit720 may output these digital values at regular intervals to thedigital-to-analog converter 742, such that the digital-to-analogconverter may generate an analog representation of thenon-zero-magnitude signal 734.

The lighting control device 700 comprises a controllable switch 744,which receives the non-zero-magnitude signal 734 and is controlled bythe control circuit 720 to generate the combined signal 730.

The lighting control device 700 also comprise a filter circuit 746 forfiltering the combined signal 730 to generate a filtered signal V_(F)and a comparator circuit 748 for comparing the filtered signal to thereference voltage V_(REF) to generate a zero-crossing signal V_(ZC)representative of the zero-crossings of the AC power source. The controlcircuit 720 may control the controllable switch 744 in concert with thegeneration of drive signal V_(DR) to control the intensity of thelighting load to the desired dimming level (e.g., firing thecontrollable switch 708 at the same time). In an example operation, thecontrol circuit 720 may detect a zero-crossing in the signal receivedfrom the filter circuit 746 and the comparator circuit 748. The controlcircuit 720 may time the generation of a control signal for dimmingoperation based on a desired light level for a first duration of time.And, when generating the control signal to effect dimming operation, thecontrol circuit 720 may close the controllable switch 744 for a secondduration of time (e.g., the remaining time in the AC half-cycle) untilanother zero-crossing is detected. In an example, the combined signal730 may include, in the first duration of time, a signal representingthe dimmer-voltage waveform developed across a controllable conductivedevice (not shown) and, in the second duration of time, thenon-zero-magnitude signal 734 from the digital-to-analog converter 742.

The combined signal 730 may be used for purposes alternative to or inaddition to serving as input to the filter circuit 746. A lightingcontrol device may provide the combine signal 730 to a processingcircuit (not shown) for performing other operations. For example, thecombined signal 730 may be provided to the processing circuit inaddition to the filter circuit 746. Also for example, the combinedsignal 730 may be provided to the processing circuit, and the filtercircuit 746 and comparator circuit 748 may be omitted.

The processing circuit may include the control circuit 720 and/or anyother device, system, and/or subsystem suitable for processing a signal(e.g., measuring, analyzing, transmitting, conveying, multiplexing,combining, modulating, and/or otherwise performing operation(s) onand/or on the basis of the signal). For example, the processing circuitmay use the combined signal 730 to perform measurements and/orcalculations regarding the AC voltage and/or current waveforms. Forexample, the processing circuit may use the combined signal 730 toperform a power calculation. For example, the processing circuit may usethe combined signal 730 in connection with circuits that benefit from afull AC voltage waveform to operate, such as circuits that benefit froma full AC voltage waveform for timing information, for example.

FIG. 8 is a simplified flowchart of a zero-crossing procedure 800executed periodically by a control circuit (e.g., the control circuit720 of the lighting control device 700) at step 810 (e.g., in responseto the zero-crossing signal V_(ZC)). First, the control circuit 720drives the drive signal V_(DR) to the gate drive circuit 714 low (e.g.,to approximately circuit common) at step 812, such that the FETs 710,712 are non-conductive at the beginning of the half-cycle. At step 814,the control circuit 720 controls the controllable switch 744 to be open(i.e., non-conductive), such that the combined signal 730 is equal tothe dimmer-voltage signal 732. The control circuit 720 then recalls thefiring time T_(FIR) (i.e., the firing angle) for the present half-cycleto control the intensity of the lighting load to the desired dimminglevel at step 816. At step 818, the control circuit 720 loads the firingtime T_(FIR) into timer A and starts timer A, decreasing in value withrespect to time, before the zero-crossing procedure 800 exits. Thecontrol circuit 720 will execute a dimming timer procedure 900 whentimer A expires.

FIG. 9 is a simplified flowchart of the dimming timer procedure 900,which is executed by the control circuit 720 when timer A expires atstep 910 (i.e., after the firing time T_(FIR) from the beginning of thehalf-cycle). The control circuit 720 drives the drive signal V_(DR) high(e.g., to approximately the DC supply voltage V_(CC)) at step 912 torender the FETs 710, 712 conductive, and then closes the controllableswitch 744 at step 914, such that the digital-to-analog converter 742 iscoupled to the filter circuit 746. Since the magnitude of thedimmer-voltage signal 732 decreases to approximately zero volts when theFETs 710, 712 are rendered conductive, the combined signal 730 is equalto the non-zero-magnitude signal 734 generated by the digital-to-analogconverter 742 after the firing time T_(FIR). The dimming timer procedure900 procedure then exits, and the zero-crossing procedure 800 will beexecuted at the next zero-crossing as determined by the zero-crossingsignal V_(ZC).

FIG. 10A is a simplified schematic diagram of a portion of a two-wirelighting control device 1000, which uses pulse-width modulation togenerate a non-zero-magnitude signal. FIGS. 10B and 10C depictcorresponding signal diagrams. As in the lighting control device 700 ofFIG. 7, the lighting control device 1000 comprises two FETs 1010, 1012in anti-series connection as the controllably conductive device adaptedcoupled in series between the AC power source and the lighting load. TheFETs 1010, 1012 are rendered conductive and non-conductive in responseto a drive signal V_(DR) provided to a gate drive circuit 1014 by acontrol circuit 1020 to control the amount of power delivered to thelighting load.

The lighting control device 1000 comprises a multiplexer 1040 forgenerating a combined signal 1030, a filter 1046 for filtering thecombined signal, and a comparator circuit 1048 for generating azero-crossing signal representative of the zero-crossings of the ACpower source. The multiplexer 1040 receives a first signal 1032, whichis received from the controllably conductive device, for example, at anormally closed (NC) input of the multiplexer. The first signal 1032 isrepresentative of the dimmer-voltage waveform developed across thecontrollably conductive device. The lighting control device 1000comprises a first resistive divider including resistors R1050, R1052,which are coupled in series between the hot terminal H and circuitcommon, and have resistances of, for example, approximately 784 kΩ and15Ω, respectively. The lighting control device 1000 also comprises asecond resistive divider including resistors R1054, R1056, which arecoupled in series between the dimmed-hot terminal DH and circuit common,and have resistances of, for example, approximately 784 kΩ and 15 kΩ,respectively.

During the positive half-cycles, current is conducted from the hotterminal H and through the first resistive divider and the body diode ofthe second FET 1012, such that a scaled voltage representative of thedimmer-voltage waveform across the controllably conductive device isgenerated by the first resistive divider. During the negativehalf-cycles, current is conducted from the dimmed-hot terminal DH andthrough the second resistive divider and the body diode of the first FET1010, such that a scaled voltage representative of the dimmer-voltagewaveform across the controllably conductive device is generated by thesecond resistive divider. The junction of the resistors R1050, R1052 ofthe first resistive divider is coupled to the reference voltage V_(REF)through a resistor R1058 (e.g., having a resistance of approximately5.49 kΩ), and the junction of the resistors R1054, R1056 of the secondresistive divider is coupled to the reference voltage V_(REF) through aresistor R1059 (e.g., having a resistance of approximately 5.49 kΩ).Accordingly, the scaled voltages generated by the first and secondresistive dividers are referenced about the reference voltage V_(REF)(and not referenced to circuit common).

The first signal 1032 is generated by combining the scaled voltagesgenerated by the first and second resistive dividers. However, since thescaled voltages generated by the first and second resistive dividers arerectified by the body diodes of the FETs 1010, 1012, the output of thefirst resistive divider is coupled to the output of the second resistivedivider by an inverting circuit 1060, which comprises an operationalamplifier (“op-amp”) 1062. The scaled voltage generated by the firstresistive divider is coupled to the non-inverting input of the op-amp1062 via a resistor R1064 (e.g., having a resistance of approximately464 kΩ). The non-inverting input is coupled to the output of the op-amp1062 via a resistor R1066 (e.g., having a resistance of approximately464 kΩ), and the inverting input of the op-amp is coupled to thereference voltage V_(REF). The output of the op-amp 1062 is coupled tothe output of the second resistive divider via a resistor R1068 (e.g.,having a resistance of approximately 5.49 kΩ). Accordingly, the firstsignal 1032 looks like an AC voltage waveform that is reference aboutthe reference voltage V_(REF), such that the magnitude of the firstsignal is always positive. In other words, the first signal 1032 isrepresentative of the dimmer-voltage waveform generated across thecontrollably conductive device (which is an AC voltage waveform), butdoes not drop below zero volts, such that the first signal may beprocessed by standard digital circuitry.

The non-zero-magnitude signal comprises a step-sine wave 1034 generatedby a pair of complementary pulse-width modulated channels on the controlcircuit 1020: a V_(PWM_LO_SINE) channel 1070 and a V_(PWM_HI_SINE)channel 1072. The functionality of the pulse-width modulated channelsmay be an available feature of a microprocessor-based control circuit.The pulse-width modulated channels may be pulse-width modulated channelsavailable on a microprocessor, a dedicated integrated circuit, composedof fundamental circuit elements, and the like. The complementarypulse-width modulated channels output of the control circuit 1020 viarespective resistors R1074, R1076 and respective capacitors R1078,R1080. The pulse-width modulated channels output across a storage RCcircuit, including storage capacitor C1082 and corresponding resistorR1084 relative to the filter reference voltage V_(REF). The step-sinewave 1034 generated by the complementary pulse-width modulated channelsand accompanying circuitry is provided to a normally-open (NO) input ofthe multiplexer 1040.

The multiplexer 1040 may switch between the NC input (i.e., the firstsignal 1032) and the NO input (i.e., the step-sine wave 1034) inresponse to a select-input control signal V_(MUX) provided at a MuxControl output of the control circuit 1020. The select-input controlsignal V_(MUX) is coupled to a select input (IN) of the multiplexer 1040via a circuit comprising a transistor Q1086 and a resistor R1088. Thecontrol circuit 1020 may signal at the Mux Control output in concertwith controlling the dimming operation of the FETs 1010, 1012 such thatthe first signal 1032 and the step-sine wave 1034 are combined to formthe combined signal 1030 at the output COM pin of the multiplexer 1040.When the select-input control signal V_(MUX) is driven high (i.e., toapproximately the DC supply voltage V_(CC)), the transistor Q1086 isrendered conductive and the select input IN of the multiplexer 1040 ispulled down to circuit common, such that the first signal 1032 at the NCinput is provided at the output COM pin of the multiplexer. When theselect-input control signal V_(MUX) is driven low (i.e., toapproximately circuit common), the transistor Q1086 becomesnon-conductive and the select input IN of the multiplexer 1040 is pulledup towards the DC supply voltage V_(CC) through the resistor 1088, suchthat the step-sine wave 1034 at the NO input is provided at the outputCOM pin of the multiplexer.

In operation, the complementary pulse-width modulated channelsVP_(PWM_LO_SINE), V_(PWM_HI_SINE) add or subtract charge to and from thestorage capacitor C1082 to adjust the magnitude of the step-sine wave1034 with respect to the reference voltage V_(REF). As illustrated inFIG. 10B, in each positive half-cycle of the step-sine wave 1034, thePWM_LO_SINE channel 1070 stays in a high default state, while thePWM_HI_SINE channel 1072 is controlled through a series of pulses to addcharge to the storage capacitor C1082 when the PWM_HI_SINE channel 1072is high and to subtract charge from the storage capacitor C1082 when thePWM_HI_SINE channel 1072 is low. By controlling the duty cycle of thepulses of the PWM_HI_SINE channel 1072 (i.e., the amount of time thatthe PWM_HI_SINE channel 1072 is high versus low), the control circuit1020 is able to adjust the magnitude of the step-sine wave 1034 in eachpositive half-cycle. In each negative half-cycle of the step-sine wave1034 as illustrated in FIG. 10C, the PWM_HI_SINE channel 1072 stays in alow default state, while the PWM_LO_SINE channel 1070 is controlledthrough a series of pulses to subtracts charge from the storagecapacitor C1082 when the PWM_LO_SINE channel 1070 is low and to addcharge to the storage capacitor C1082 when the PWM_LO_SINE channel 1070is high.

As illustrated, the width of the pulse may affect the rate of change ofthe step-sine wave 1034. For example, changing the width of each pulseof the PWM_HI_SINE channel 1072 may change the overall rate of change inthe step-sine wave 1034 in the positive direction. Similarly, changingthe width of each pulse of the PWM_LO_SINE channel 1070 may change theoverall rate of change in the step-sine wave 1034 in the negativedirection.

To generate the appropriate duty cycles for the pulse-width modulatedchannels, the control circuit 1020 may have stored one or more tables.The control circuit 1020 may control the state of each of thepulse-width modulated channels, performing a “SET” when a timing counteris zero and a “RESET” when the counter reaches a value, such as a valuestored in a register. In light of the SET/RESET structure, the frequencyof the step-sine wave 1034 may be configurable, for example, to matchthe line frequency. An AC half-cycle may be divided into a number ofbins, and the modulo of a timer channel may be used to control theperiod of the half-cycle. The modulo may be 1/N_(BINS) of the half-cycleperiod where N_(BINS) is the number of bins used. The timer channel maycalculate the half-cycle period on a half-cycle-by-half-cycle basis.Also, for example, the number N_(BINS) of bins in the half-cycle may beselected to be an exponent of two. To illustrate, the number N_(BINS) ofbins may be 32 per half-cycle to provide acceptable resolution andpreserve processing capacity by limiting the number of interruptsrequired.

One or more lookup tables may be created to generate the step-sine wave1034. Each value from the lookup table may be fed into the valueregister. At each timer overflow, the next value may be loaded. Eachvalue, when loaded, may establish the voltage value for that bin. Ineffect, the voltage step in each bin may be proportional to the dutycycle in that bin, and the duty cycle is provided by the value from theregister. The frequency of the step-sine wave 1034 may be set based on alook-up table, created from a sine table. Here, the duty cycle is“value/modulo,” and the “modulo” is based on the line frequency (e.g.,modulo=f/2*32, for 32 bins).

The table values may be scaled. The scaled tables may have athree-element depth, which may allow read and write functions each tohave ownership of one element, with one element in redundancy. The taskexecution may be asynchronous, and a page switching scheme may be usedto make sure that reading and writing avoid overlapping. Also, the pageswitching scheme may be used to ensure that table updates are absorbedat the zero cross.

In operation, the control circuit 1020 may have a number of interrupts.At each interrupt, two functions may be performed. The first functionmay include loading a new value for the next bin. This action may beperformed every time the interrupt occurs. The second function mayinclude prepping the timer channels for the next half cycle. Thisfunction may be performed at the zero-cross (i.e., starting bin zero),when the modulo may be updated to match the latest value for theupcoming half cycle. At this point, the scaled sine table may also bechecked for a new valid page.

The register buffers may require management. In some microprocessorsthere may be an inherent delay built into the pulse-width modulationregisters. The delay may be caused by registers being buffered where thebuffer value is applied to the timer channel when the timer counter isreset. The buffering ensures that a single pulse-width modulation cycleis completed before changing the parameters. To account for thisbuffering, loading a value for a given bin may be performed one bin inadvance.

Processing each zero-crossing may introduce latency into the system. Forexample, there may be a certain amount of delay attributed to thezero-cross interrupt service routine. Phase error may be calculated andcorrected in the step-sine signal. For example, the phase error may becalculated based on difference of a real zero-cross and the zero-crossof the step-sine wave 1034. The phase error may be calculated by thecontrol circuit 1020 before loading the modulo for the next half-cycle.Once the phase error is calculated, one or more correction values may beestablished. The one or more correction values may be selected such thatthe phase error will be canceled out by the next zero-cross. Forexample, a correction value equal to the phase error divided by thenumber of bins may be added to each bin for the subsequent half-cycle.With this correction in the subsequent half-cycle, the step-sine wavewill “catch up” to the real zero crossing. With continuous correction,the step-sine wave may maintain synchronicity with the AC source signal.

FIG. 11 is a simplified flowchart of a bin set procedure 1100 executedperiodically by a control circuit (e.g., the control circuit 1020 of thelighting control device 1000) at the beginning of each bin of thestep-sine wave 1034 at step 1110 (e.g., once every 3.75 μsec for a 60-HzAC line voltage). The control circuit 1020 uses a variable n to keeptrack of the present bin during the half-cycle. If the variable n isequal to one at step 1112 (i.e., it is the first bin of the half-cycle),the control circuit 1020 determines the period T_(BIN) of the bins forthe present half-cycle at step 1114. For example, the control circuit1020 may calculate the period T_(BIN) using the number N_(BINS) of binsin the half-cycle and a period T_(HC) of the half-cycle, i.e.:T_(BIN)=1/N_(BINS)·T_(HC).

The control circuit 1020 may also update the value used for the periodT_(HC) of the half-cycle in response to the zero-crossings determinedfrom the zero-crossing signal V_(ZC) to account for errors or changes inthe frequency of the AC line voltage.

Next, the control circuit 1020 recalls the duty cycle DC for bin n asstored in the memory at step 1116, and determines the period T_(PLS) ofthe pulse for the present bin at step 1118, i.e.,T _(PLS) =DC·T _(BIN)At step 1120, the control circuit 720 loads the period T_(PLS) of thepulse into timer B and starts timer B decreasing in value with respectto time, such that a bin reset procedure 1200 (which will be describedin greater detail below with reference to FIG. 12) will be executed whentimer B expires. If the present half-cycle is a positive half-cycle atstep 1122, the control circuit 1020 drives the PWM_LO_SINE channel 1070high at step 1124 and the PWM_HI_SINE channel 1072 high at step 1126. Ifthe present half-cycle is a negative half-cycle at step 1122, thecontrol circuit 1020 drives the PWM_HI_SINE channel 1072 low at step1128 and the PWM_LO_SINE channel 1070 low at step 1130.

If the variable n is not equal to the number N_(BINS) of bins in thehalf-cycle at step 1132, the control circuit 1020 increments thevariable n by one at step 1134, and the bin set procedure 1100 exits. Ifthe variable n is equal to the number N_(BINS) of bins in the half-cycleat step 1132 (i.e., it is the end of the present half-cycle), thecontrol circuit 1020 sets the variable n equal to one at step 1136. Ifthe present half-cycle is a positive half-cycle at step 1138, thecontrol circuit 1020 sets the present half-cycle to negative at step1138, and the bin set procedure 1100 exits. If the present half-cycle isa negative half-cycle at step 1138, the control circuit 1020 sets thepresent half-cycle to positive at step 1140, and the bin set procedure1100 exits.

FIG. 12 is a simplified flowchart of the bin reset procedure 1200, whichis executed by a control circuit the control circuit 1020 when timer Bexpires at step 1210 (i.e., after the period T_(PLS) of the pulse). Ifthe present half-cycle is a positive half-cycle at step 1212, thecontrol circuit 1020 drives the PWM_HI_SINE channel 1072 low at step1214 and the bin reset procedure 1200 exits. If the present half-cycleis a negative half-cycle at step 1212, the control circuit 1020 drivesthe PWM_LO_SINE channel 1070 high at step 1216, before the bin resetprocedure 1200 exits. The control circuit 1020 will execute the bin setprocedure 1100 again at the beginning of the next bin.

FIG. 13 is a simplified schematic diagram of a portion of an exampletwo-wire lighting control device 1300. The lighting control device 1300comprises a control circuit 1320 that simply receives the first signal1032 that is representative of the dimmer-voltage waveform developedacross the controllably conductive device (i.e., the FETs 1010, 1012).The control circuit 1320 is operable to generate the drive signal V_(DR)for rendering the FETs 1010, 1012 in response to simply the first signal1032. The control circuit 1320 is operable to generate a digitalcombined signal S_(COMB), which is filtered using a digital Besselfilter. The control circuit 1320 uses a filtered signal S_(FILT), whichis the output of the digital filter, to determine the zero-crossings ofthe AC voltage waveform.

FIG. 14 is a simplified schematic diagram of a sampling procedure 1400that is executed periodically (e.g., at a sampling rate) by a controlcircuit (e.g., the control circuit 1320 of the lighting control device1300) in order to sample and process the first signal 1032. The controlcircuit 1320 uses a timer to keep track of the present time during eachhalf-cycle. The timer increases in value with respect to time and isreset at the beginning of each half-cycle. The control circuit 1320 isoperable to render the controllably conductive device conductive when avalue t_(TIMER) of the timer reaches the firing time T_(FIR) that isstored in memory. Referring to FIG. 14, if the value t_(TIMER) of thetimer is less than the firing time T_(FIR) at step 1410, the controlcircuit 1320 samples the first signal 1032 at step 1412 to generate asample S_(DV) that is representative of the instantaneous value of thedimmer-voltage waveform across the controllably conductive device. Thecontrol circuit 1320 then stores the sample S_(DV) as the next value ofthe digital combined signals S_(COMB)[i] at step 1414. If the valuet_(TIMER) of the timer is less than the firing time T_(FIR) at step1410, the control circuit 1320 determines the present value of a fillsignal S_(FILL) at step 1416, for example, using a lookup table, andthen stores the present value of the fill signal S_(FILL) as the nextvalue of the digital combined signals S_(COMB)[i] at step 1418. Thecontrol circuit 1320 then increments the variable i by one at step 1420and executes the digital Bessel filter on the digital combined signalS_(COMB) at step 1422. If the filtered signal S_(FILT) is not less thana signal threshold S_(TH) at step 1424, the sampling procedure 1400simply exits. However, if the filtered signal S_(FILT) is less than thesignal threshold S_(TH) at step 1424 (i.e., indicating a zero-crossing),the control circuit 1320 renders the FETs 1010, 1012 non-conductive atstep 1426 and resets the value t_(TIMER) of the timer to zero seconds atstep 1428, before the sampling procedure 1400 exits.

What is claimed is:
 1. A lighting control device comprising: a controllably conductive device for supplying an AC line voltage to a load in response to a drive signal, wherein the controllably conductive device is non-conductive for a first duration of time and is conductive for a second duration of time, wherein the first duration of time and the second duration of time are within the same half-cycle of the AC line voltage; a control circuit configured to: receive a signal from the controllably conductive device that represents a voltage developed across the controllably conductive device during the first duration of time; generate a sine-wave-shaped signal that complements the voltage developed across the controllably conductive device during the second duration of time; and filter the signal from the controllably conductive device during the first duration of time and the sine-wave-shaped signal during the second duration of time.
 2. The lighting control device of claim 1, wherein the control circuit comprises: a controller configured to generate the sine-wave-shaped signal; a combiner circuit for combining a signal from the controllably conductive device during the first duration of time and the sine-wave-shaped signal during the second duration of time; and a filter circuit configured to filter the combined signal from the combiner circuit.
 3. The lighting control device of claim 2, wherein the sine-wave-shaped signal is a step-sine wave.
 4. The lighting control device of claim 3, wherein the step-sine wave comprises steps based on at least one pulse-width modulated signal.
 5. The lighting control device of claim 3, wherein the step-sine wave comprises steps based on at least one phase correction corresponding to a zero-crossing detected by a zero crossing detector.
 6. The lighting control device of claim 2, wherein the combiner circuit comprises a switch configured to switch between the signal from the controllably conductive device and the sine-wave-shaped signal based on the drive signal, the switch configured to select the signal from the controllably conductive device during the first duration of time and the sine-wave-shaped signal during the second duration of time.
 7. The lighting control device of claim 6, wherein the switch comprises a multiplexer that switches in concert with the drive signal.
 8. The lighting control device of claim 2, wherein the filter circuit is a low-pass filter.
 9. The lighting control device of claim 8, wherein the low-pass filter provides a substantially linear phase delay of less than one-half of a period of the fundamental frequency and attenuates frequency components of a filter-input signal that are substantially equal to third order harmonics and greater of the fundamental frequency.
 10. The lighting control device of claim 1, wherein the control circuit comprises a controller configured to sample the signal from the controllably conductive device during the first duration of time.
 11. The lighting control device of claim 10, wherein the controller is configured to filter the signal from the controllably conductive device and the sine-wave-shaped signal using a digital Bessel filter.
 12. The lighting control device of claim 1, wherein the control circuit is configured to: generate a filtered signal by filtering the signal from the controllably conductive device during the first duration of time and filtering the sine-wave-shaped signal during the second duration of time; and determine a zero-crossing of the AC line voltage from the filtered signal.
 13. The lighting control device of claim 12, wherein the control circuit is configured to generate the drive signal to control the controllably conductive device based on the determined zero-crossing of the AC line voltage.
 14. The lighting control device of claim 1, wherein the sine-wave-shaped signal is shaped such that delay associated with filtering the signal from the controllably conductive device during the first duration of time and the sine-wave-shaped signal during the second duration of time is consistent with delay associated with filtering the AC line voltage.
 15. The lighting control device of claim 1, wherein the sine-wave-shaped signal is shaped to maintain stability of the two-wire lighting control device. 